High speed modem for a communication network

ABSTRACT

The present invention relates to a receiver at the head-end or a centralizing unit side in a communications system or network for signals in the upstream direction which is the direction from user to head-end or a centralizing unit that is linked to a number of users, the number being equal to or larger than one.  
     The receiver of the invention is suited for the reception of burst-mode signals. The receiver performs a channel estimation on a per-burst basis in real time or essentially immediate. The channel estimation is necessary to do successful data detection of modulated data.  
     The receiver of the invention performs the channel estimation and data detection in one compact all-digital mechanism that has no tuning parts. The reception method works in an aspect according to the principle of a matched filter receiver, but stores no local copy of the required matched waveform. Rather, a copy of the matched waveform is included in the preamble of the signals.

CONTINUING APPLICATION

[0001] This application is a continuation of U.S. application Ser. No.09/100,304, filed on Jun. 19, 1998.

FIELD OF THE INVENTION

[0002] The present invention relates to a receiver at the head-end orside in communications system or network. More specifically, theinvention relates to a receiver for signals in the upstream directionwhich is the direction from user to head-end or a centralizing unit thatis linked to a number of users, the number being equal to or larger thanone.

BACKGROUND OF THE INVENTION

[0003] The technology area of communications systems is subject to avast engineering effort in order to allow for an always increasingnumber of applications.

[0004] Currently, the public access network for television (CATV) isbeing prepared for bi-directional communication. The goal of upgradingthe CATV network is to provide bi-directional communication of digitaldata at speeds well beyond that of traditional data communication overtelephone lines. This way, CATV data communication allows new types ofapplications such as video-on-demand and fast Internet access. It alsoprovides an alternative to existing telephone services. In that case,the analog voice data is digitized and transmitted as a collection ofdata packets. For CATV communications, one defines the downstreamdirection, going from the broadcasting head-end side to the user side,and the upstream direction, going from the user side to thebroad-casting head-end side. These definitions of upstream anddownstream direction in the sequel are adopted also for anycommunications system or communications network. The upstream directionis defined as going from a user side or a subscriber residence to acentral office or head-end or a centralizing unit that is linked to anumber of users, the number being larger than one.

[0005] The communication process between the end-user and head-end istypically organized as a number of hierarchical subprocesses, eachrunning at a different level of abstraction. This hierarchy is needed toexpress the access of the communications network between communicatingparties in an efficient way. This invention is concerned with thecommunication subprocess on the lowest hierarchical layer, i.e., thephysical layer.

[0006] The goal of communications on the physical layer is:

[0007] to provide a reliable means of data communication by applyingmethods of data modulation and demodulation, and

[0008] to transfer this reliable communication method towards the higherlayers. This is done by means of an interface and a protocol.

[0009] The topology of the physical layer a typical public accessnetwork for television is shown in FIG. 1. Both the head-end side anduser side have a transmitter and receiver to make bi-directionalcommunication possible. The network has a tree-like topology, andconsists of both active elements (bi-directional amplifiers) and passiveelements (cable, splitters and taps). For each head-endtransmitter/receiver pair, many user-end transmitter/receiver pairs mayexist. Typically, 400 users can be served through 1 head-end. The up-and downstream communications path run over a single electrical path andare differentiated through frequency multiplexing, as shown in FIG. 2.

[0010] U.S. Pat. No. 3,962,637 and U.S. Pat. No. 5,127,051 describeimproved modems for high frequency data transmission. In particular,U.S. Pat. No. 5,127,051 discloses a modem system that can adapt to fastchannel variations by rapidly deriving accurate channel estimate withoutexcessive storage of data overhead. Accordingly, the data aretransmitted in a frame carrying at least two identical data sequences.This approach, however, has as a disadvantage a major overhead forstatic type of channel and therefore has a slow performance for thesechannel types. The publication “CATV Return Path Characterization forReliable Communications” by C. A. Eldering, IEEE CommunicationsMagazine, August 1995 addresses the problem of reliable solutions forbi-directional communication. In said publication, an emphasis is givento the key problem of the understanding of the communication channelcharacteristics in the upstream direction. One of the problems inupstream communication is as follows. As we move from the user to thehead-end, the physical transmission medium (the cable or the channel) isshared by an increasing amount of users. Therefore, users will share themedium in the upstream direction by means of an appropriate multi-accessprotocol. This invention is concerned with the time-divisionmulti-access protocol which uses burst-mode signals. In this protocol,each user gets in turn connection to the head-end during a fixedtime-slot. The start and end of the time-slot is decided at the head endby allocation algorithms running in the higher hierarchicalcommunication layers. A discussion of these allocation algorithms is outof the scope of this patent application since it is concerned with thetransport of data on the physical layer only.

[0011] The key problems to solve in order to establish a reliableupstream communication between the user and the head-end are thefollowing:

[0012] As the signals emitted at the user side propagate through theupstream channel, they are attenuated and delayed. This attenuation anddelay is different for each user, since each user is connected at adifferent position in the tree network as seen from the head-end.Therefore the head-end must estimate these modulation distortions on aper-burst basis. It must also do this as' fast as possible, since duringestimation time, no useful data can be transmitted.

[0013] Besides attenuation and delay, the signals also suffer from groupdelay distortion. Group delay distortion is caused by the non-linearphase characteristics of the (mainly active) components located in theup-stream channel. The effect of group delay distortion is that thetime-domain shape of the distorted signal is changed. The distortion isa linear effect, which means that it can be removed by passing thereceived signal through a proper filter before detecting it. Therequired shape of this filter is dependent on the amount and type ofgroup delay distortion, and is again different for each user. Therefore,the head-end receiver must estimate the coefficients of this filter on aper burst basis. Failure to do so causes an effect at the head-endreceiver side called inter-symbol-interference (ISI). ISI degrades thequality of the data detection process, and therefore should be avoided.

[0014] The process of estimating attenuation, delay and group delaydistortion is jointly called channel estimation.

[0015] The upstream communications CATV path is also susceptible ofnoise influences. These can be caused by electrical appliances orspurious emissions of radio-band users (mobile communication, amateur,CB, . . . ), and other, unknown sources. Since the actual time-domainshape of noise is unknown, it cannot be removed at the receiver. It willtherefore also degrade data detection performance of the receiver. Thetransmission can however be protected against noise influences byapplying a proper encoding of the data. The encoding increases theredundancy of the transmitted data pattern. At the receiver side, theremoval of this redundancy can then be used to identify locations oferrors in the received data pattern. Eventually, the redundancy can evenbe used to correct the errored values.

[0016] All estimation processes active in the head-end receiver mustproceed as fast as possible. During the estimation the actual delay,attenuation and group delay distortion is unknown and no data can bedetected successfully. The signal transmitted by one user is of a burstynature. Therefore, the shorter the estimation time, the more time willbe left in the signal burst that can be used for the transmission ofactual data.

[0017] In the remainder of this document, we will first summarize thekey properties of the invention, which is a digital receiver for theseburst mode signals. Next, we will give a detailed description of thereceiver and its operation.

SUMMARY OF THE INVENTION

[0018] The present invention relates to a receiver at the head-end or acentralizing unit side in a communications system or network. Thereceiver is adapted for receiving signals in the upstream directionwhich is the direction from user to head-end or a centralizing unit thatis linked to a number of users, the number being equal to or larger thanone. The present invention further relates to communication systemsmaking use of burst-mode signals.

[0019] The present invention relates to a telecommunication system withmeans for upstream communication from a user to a head-end over achannel, said means for upstream communication including a receivercomprising a detect unit being configured in a feed-forward data drivenarchitecture.

[0020] Each algorithm is executed by a dedicated digital hardwaremachine, comprising a local controller and a data path. The data pathexecutes the data processing operations inside the algorithm, while thelocal controller performs operation sequencing, and algorithmsynchronization.

[0021] In an aspect of the present invention, said detect unit isadapted for receiving a burst-mode signal, said burst-mode signal havinga preamble with at least one training portion at the beginning of theburst followed by at least one timing alignment portion.

[0022] Said signal further can comprise a user message.

[0023] In another aspect of the invention, the detect unit of the systemcan comprise a block for extracting information on at least onetransmission characteristic of said burst-mode signal in said channel,said information being obtained as the coefficients of a fractionallyspaced feed-forward equalizer in said block.

[0024] The receiver can further comprise a timing block wherein saidalignment portion is processed, said alignment portion providing thetransition from said training portion to said user message as thedownsampling phase of said block.

[0025] Yet the receiver can comprise a detection block for detectingsaid signal and adjusting the power level of said signal to apredetermined power level; and a filter block with programmablecoefficients for filtering said user message.

[0026] Said coefficients can be extracted from said preamble inreal-time.

[0027] Yet in a further aspect of the invention, the receiver has afeed-forward architecture that is configured as a chain of subsequentcomponents, said signal being consecutively passed and without feedbackthrough said chain, the chain comprising components having a finitestate machine and a data path, the signal being passed through the datapaths, the finite state machines running a control program, saidcomponents behaving differently when a burst signal is received or not.

[0028] Yet the present invention is also related to, in a communicationsystem, a method for transmitting a signal, said method comprising thesteps of:

[0029] transforming said signal into a first sequence of digital data;

[0030] prepending a predetermined sequence of data to said firstsequence of data, said predetermined sequence having a training portionat the beginning of the predetermined sequence followed by a timingalignment portion, the sequence of said predetermined sequence and saidfirst sequence being a resulting data sequence; and

[0031] modulating said resulting data sequence to a predetermined formatfor transmission.

[0032] The method can further comprise the step of receiving said signalin a receiver with an equalizer block with programmable coefficients,said step comprising the substeps of:

[0033] fixing said coefficients while analyzing said training portion ofsaid predetermined sequence of first data; and

[0034] detecting said timing alignment portion as the transition to saidfirst sequence of data; and thereafter

[0035] performing data filtering on said first sequence of data.

[0036] The method can be executed in real-time.

[0037] In a further aspect of the invention, a method of operating anadaptive modem for analyzing signals being transmitted over acommunications channel is disclosed. Said signals are being sent in atleast one burst comprising a preamble and a user message, said methodcomprise the steps of:

[0038] receiving the transmitted signals;

[0039] generating a plurality of coefficients for a downsampling feedforward adaptive equalizer from a training sequence in said preamble ofsaid burst;

[0040] adapting said downsampling feed forward adaptive equalizer tosaid communications channel.

[0041] According to the method of the invention, the signals areanalyzed on a burst-by-burst base and in real-time.

[0042] The receiver of the invention is suited for the reception ofburst-mode signals. The receiver performs a channel estimation on aper-burst basis in real time or essentially immediate. The channelestimation is necessary to do successful data detection of modulateddata.

[0043] Current state of the art modems do not perform per-burst channelestimation and/or group delay distortion estimation, but rather assume afixed channel from which the data can be detected by means of a fixeddata filter.

[0044] The receiver of the invention performs the channel estimation anddata detection in one compact all-digital mechanism that has no tuningparts. The reception method works in an aspect according to theprinciple of a matched filter receiver, but stores no local copy of therequired matched waveform. Rather, a copy of the matched waveform isincluded in the preamble of the signals.

[0045] The burst-mode signals that are received comprise two parts: Apreamble and a payload. The preamble has among other functions, thefunction to perform the channel estimation and synchronize thedemodulation loops, while the payload contains the actual data totransmit, including error correcting codes. The preamble comprises atleast one training portion followed by a timing alignment portion. Incase several training portions are included in the preamble, the methodof the invention can be implemented as an averaging algorithm, averagingthe results obtained from the different training portions.

[0046] The overhead of the burst-mode signal is therefore primarilylocated in the preamble, since it contains no user data. The presentinvention contains a very short and fixed preamble.

[0047] The channel estimation method allows compensation of an arbitraryamount of group delay distortion through a very simple extension of theburst preamble.

[0048] The channel estimation is solely based on the burst preamble andis therefore very fast.

[0049] Due to the use of a training sequence, no differentialencoding/decoding of the symbols is required, as is traditionally seenon QAM type modems.

[0050] The reception method allows reception of different pulse shapeswith one single implementation. When the pulse shapes are of theso-called root-raised-cosine (RRC) family (which is the most commonlyused shape in state-of-the art quadrature modulated modems), differentRRC roll-off factors are supported by one and the same architecture.Other pulse shapes are possible as long as they have the ISI-freeproperty. This property is discussed in the sequel.

[0051] In general, the reception method implies that one and the samereceiver can be used to receive different standards (self-adapting).

[0052] It uses a feed-forward architecture. For a burst-mode receiver,this is an important property. A feed-forward architecture isself-controlled and pipelineable. Due to the absence of feedback loops,system stability is independent of the delay of individual components.

[0053] The receiver and method of the invention furthermore in otheraspects allow for an immediate channel estimation and allow tocompensate for substantially any group delay distortion.

[0054] The receiver and method of the invention in yet another aspectallow for combined timing and phase estimation. The receiver of theinvention also in an aspect is adapted for receiving signals oftransmitters having different characteristics as being for instance theproducts of different companies or having different roll-off factorssuch as determined in the DAVIC standard or the IEEE standard or theMCNS standard.

BRIEF DESCRIPTION OF THE DRAWINGS

[0055]FIG. 1 shows an example of a CATV network for data communications.

[0056]FIG. 2 shows an example of a method of Frequency Multiplexing toseparate up- and downstream directions in the CATV network of FIG. 1.

[0057]FIG. 3 shows an example of the Physical Layer of a communicationsystem with means for upstream communications.

[0058]FIG. 4 shows the characteristics of Group delay on a typicalcommunications channel.

[0059]FIG. 5 shows a Mathematical model of a preferred embodiment of acommunication system according to the present invention.

[0060]FIG. 6 shows an example of a QPSK and QAM16 constellation in anaspect of the invention.

[0061]FIG. 7 shows a graphical representation of the modulation processfor m=2.

[0062]FIG. 8 shows an ISI-free pulse for m=2.

[0063]FIG. 9 shows the signal format for the communication system of theinvention.

[0064]FIG. 10 shows an example of a Generic Receiver Architectureaccording to a preferred embodiment of the invention.

[0065]FIG. 11 shows an example of a Receiver Architecture according toan example of the invention.

[0066]FIG. 12 shows an example of a DOC block of FIG. 11.

[0067]FIG. 13 shows Spectra in the DOC (a) at the input (b) after themixer (c) at the output.

[0068]FIG. 14 shows Halfband Filter Requirements for the receiverarchitecture of FIG. 11.

[0069]FIG. 15 shows an example of a AGD block of FIG. 11.

[0070]FIG. 16 shows an example of AGD operation FIG. 11.

[0071]FIG. 17 shows the derivation of the data filter coefficients outthe preamble.

[0072]FIG. 18 shows an MFE block FIG. 11.

[0073]FIG. 19 shows a Correlator Operation.

[0074]FIG. 20 shows an LMS block FIG. 11.

[0075]FIG. 21 shows the operation of a DMP block.

[0076]FIG. 22 shows a System Implementation according to the invention.

[0077]FIG. 23 shows an Implementation using a feed-forward-FSM structureaccording to another preferred embodiment of the invention.

[0078]FIG. 24 shows Noise Performance. Circle=Theory (QAM16 or QPSK).Diamond=QPSK mode. Cross=QAM16 mode.

[0079]FIG. 25 shows an AGD operation. Dependence on Threshold.

[0080]FIG. 26 shows an example of AGD Range sensitivity.

[0081]FIG. 27 shows an amplitude distortion performance.

[0082]FIG. 28 shows an amplitude ripple model.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENT

[0083] For the purpose of teaching of the invention, several embodimentsof the invention will be described in detail. This detailed descriptionin no way however can be used to limit the present invention, the scopeof the invention being limited only by the terms of the appended claims.

[0084] Relevant characteristics of the different embodiments of theinvention are summarized below and comparison to current state-of-theart devices is given;

[0085] The data transmission proceeds in a rather narrow spectral band(approx. 5-65 MHz), compared the required data rate necessary to deliverbroadband data services (1-10 Mbit/s).

[0086] Thanks to the real time channel estimation of the receiver of theinvention, 16-ary QAM can be applied successfully.

[0087] The upstream data rate is 10 Mbps. However, since a singledigital processor is used for data detection, varying the symbol rate ismerely a matter of varying the clock frequency by which the digitalprocessor is clocked. Furthermore, the data to transmit is quadraturedemodulated to obtain the highest spectral density. This spectraldensity optimization allows efficient use of the upstream spectralbandwidth. QPSK (quadrature phase-shift keying) modulation is thetechnique of choice for current state of the art cable modems.

[0088] The invention presented supports QPSK, but also the spectrallymore efficient 16-ary QAM (quadrature amplitude modulation). It is to benoted that the receiver of the invention can also be used for dataspeeds of 1 Mbit/s or lower and 100 Mbit/s or 1 Gbit/s or higher.

[0089] The receiver delivers a symbol error rate of 5.10⁻⁵ at a carrierto noise power level of C/N=23 dB for QAM16. The theoretical optimalreceiver for QAM16 offers the error rate at a carrier to noise powerlevel of C/N=19.5 dB under the condition of white gaussian noise.

[0090] The receiver is a complete, digital single-chip receiver. Analogdata is fed in by means of a fixed-clock analog-to-digital converter.

[0091] Some numerical properties of the receiver architecture aresummarized in table 1.

[0092] In the following descriptions, preferred embodiments of theinvention are described. First, an overview of the general communicationsystems is given. Next, the receiver requirements are extracted out of amathematical system model. Then, the receiver and its operation will bediscussed according to preferred embodiments of the invention. TALBE 1Upstream Receiver Parameters of the embodiments of the inventiondescribed in the sequel Parameter Value Unity Symbol rate 2.5 Msym/s Bitrate 10 Mbit/s (QAM16) Modulation QAM16 or QPSK Sample frequency 10 MHzMulti-access Scheme TDMA Preamble length 17 Symbols

[0093] These values are only for orientation and sample systemdescriptions.

PREFERRED EMBODIMENTS OF THE INVENTION

[0094] Communications System Overview

[0095]FIG. 3 shows an overview of an upstream CATV communication systemthat resides between the user (1) and the head-end (2). In this Figure,abstraction was made from all communication hierarchy above the physicallayer.

[0096] The following parts are discerned:

[0097] A transmitter (3). The transmitter (3) reads a user message, andmodulates this into the burst-format required for transmission over theupstream CATV channel (4).

[0098] The transmitter comprises a source coder (5), a channel coder(6), a preamble insertion mechanism (7), and a modulator (8). The sourcecoder (5) translates the user message from a specified input format(analog voice data, images, . . . ) to a sequence of digital data. Thechannel coder (6) increases the redundancy present in the sequence tomake the transmission more robust to noise influences on the channel.The preamble insertion unit (7) prepends a known sequence of data ontothe burst data sequence. As indicated above, the purpose of the preambleis to speed up the channel estimation process in the receiver (9).

[0099] Finally the modulator (8) receives the resulting data sequenceand converts it to a format suitable for upstream CATV transmission.This includes conversion of the discrete sequence to a continuous shape,and frequency translation of the modulated signal to the desired carrierfrequency. For upstream CATV communication, this frequency is chosenbetween 5 and 65 MHz.

[0100] A channel model. Rather then representing the actual channel, amodel is given of the distortions occurring in the channel. The delay,attenuation and group delay distortion all are linear distortions thatcan be modelled by means of a filter (10).

[0101] As an example, the group delay as a function of frequency for thetypical CATV upstream path is shown in FIG. 4. The group delay value isdefined as the differential of the channel phase characteristic withrespect to frequency. It can be understood as the time difference ittakes for signal components at a different frequency to travel over thechannel.

[0102] The modulated signal, which extends over a certain bandwidth,gives rise to group delay distortion. The distortion is merely an effectof the non-linear phase characteristic of the channel (4). In this case,a filter (10) is provided in the system model which exhibits the samephase characteristic as the channel. The receiver (9) then introduces aninverse phase distortion according to the compensating characteristicshown in the FIG. 4.

[0103] The noise is modelled by introducing a random signal (11) withunknown shape but known probability distribution. The distributionvaries with the type of disturbance that has to be modelled. For a largenumber of unknown disturbances, a gaussian probability is used. Theinfluence of the noise on the signal is expressed as the relative powerlevel of the desired signal to the noise signal. This is also calledsignal-to-noise ratio.

[0104] A receiver (9). The receiver (9) accepts the burst signaltransmitted by the transmitter (3), performs channel estimation, anddetects the data symbols that were inserted in the transmitter modulator(8). All of these occur in the Detect block (12). After detection,channel decoder (14) performs error correction using the redundancyintroduced earlier, while source decoder (15) translates a discretesequence back to the desired source format.

Derivation of Receiver Requirements

[0105] In this section, the requirements for the Detect block (12) ofthe receiver (9).

[0106]FIG. 5 shows a transmitter (3), a channel (4) model and a receiver(9) are mathematically formulated. For the transmitter and receiver, thesource coding and channel coding have been omitted, and only the innerparts are shown. The model is formulated by means of a Z-transform. Itis also shown in baseband-equivalent format, which means that thefrequency translations have been omitted (the up-conversion at thetransmitter side and the down-conversion at the receiver side). Thisrequires the channel model to be expressed in baseband equivalent formattoo.

[0107] In order to represent a quadrature modulated signal in basebandequivalent format, complex values are needed. The fat arrows in thefigure represent complex signals, while the thin ones (21) (22)represent real signals.

[0108] The sequence of operations performed in this system is asfollows. The transmitter (3) reads in a data sequence V,(z) and mapsthis to a QPSK or QAM-16 symbol sequence S_(t)(z). A QPSK symbol cantake on 4 different (complex) values, while a QAM16 can take on 16.Since these signals are complex values, it is easy to represent them ina two-dimensional plane with the real component along the X-axis and theimaginary component along the Y-axis. Mapping the 4 (QPSK) or 16 (QAM16)different symbol values onto this plane, we arrive at the constellationplots, as shown in FIG. 6. The real component is indicated with thesymbol I (for in-phase), while the imaginary component is shown withsymbol Q (for quadrature).

[0109] After mapping of the data sequence V_(t)(z), the symbol sequenceS_(t)(z) is obtained. The symbol sequence S_(t)(z) is then up-sampled bya factor m and fed into the pulse shaping filter T(z). The equivalentbaseband transmitted signal thus is given by:

S _(c)(z)=S _(t)(z ^(m))T(z)  (1)

[0110] A graphical representation of this modulation process is shown inFIG. 7 for the case of m=2. In this situation the digital modulator issaid to operate at two samples per symbol.

[0111] In the channel (4) model, the following effects are successivelyapplied on the signal.

[0112] A frequency and/or phase error can be introduced by means of acarrier signal F(z) and a mixer. The frequency/phase errors are causedby a mismatch of the frequency up-conversion in the transmitter (3) andthe frequency down-conversion in the receiver (9). At the receiver, theobserved signal then is misaligned in terms of carrier frequency andphase. The signal F(z) represents this misalignment.

[0113] For a constant phase error Φ

F(z)=Z(e ^(−j.Φ))  (2)

[0114] and for a frequency error γ

F(z)=Z(e ^(j.2.pi.t.γ)))  (3)

[0115] with Z( ) denoting the Z-transform operation.

[0116] The delay, attenuation and group delay distortion is added insidea filter C(z).

[0117] Noise N(z) is added to the signal.

[0118] The resulting received signal S_(d)(z) thus is given by:   (4)

[0119] f_(s) is the sample frequency of the discrete system. In thereceiver, this signal is filtered by R(z), and down-samples back tosymbol rate. The resulting sequence S_(r)(z) can then be inverse-mappedto the received data sequence V_(r)(z).

[0120] We next derive the requirements for the filter R(z) for differenttypes of distortion.

[0121] Receiver Requirements for an Ideal Channel

[0122] In case of an ideal channel, the incoming signal in the receiver(9) is given by:

S _(d)(z)=S _(c)(z)=S_(t)(z ^(m))T(z)  (5)

[0123] After filtering by R(z) we find:

S _(r)(z ^(m))=S _(t)(z ^(m))T(z)R(z)  (6)

[0124] For correct operation, T(z)R(z) is required to have the ISI-freeproperty. The property of ISI-freeness requires two conditions to befulfilled for T(z)R(z):

[0125] At m samples per symbol the T(z)R(z) is to be zero for allz-powers that are multiples of m.

[0126] For z-power zero, T(z)R(z) is also required to be one.

[0127] Provided this ISI-free condition is fulfilled, the downsamplingoperation yields:

S_(r)(z)=S_(t)(z)  (7)

[0128] Thus, the transmitted symbol sequence is identical to thereceived symbol sequence.

[0129] The property of ISI-freeness for R(z)T(z) is representedgraphically in FIG. 8 for the case m=2.

[0130] In QPSK and QAM16 modulation, the ISI-free pulse shape T(z)R(z)is commonly a raised cosine pulse (RC).

[0131] Besides the ISI-free condition for T(z) and R(z), an additionalcondition is used to determine the individual contributions of bothfilters to the RC pulse. This condition is the matched filterrequirement. It can be shown that a matched filter (a filter whose timedomain impulse response is the reverse of a given filter impulseresponse) maximizes the signal-to-noise ratio of the data detection, andhence minimizes errors.

[0132] Therefore, ISI-free operation requires T(z) and R(z) to be equaland matched:

R(z)=T*(z ⁻¹)  (8)

[0133] The * notation indicates complex conjugation. This required pulseshape is called a square-root raised cosine. Because this is asymmetrical pulse, original and matched shapes are identical.

[0134] We summarize that a square-root raised cosine pulse is therequired shape for both T(z) and R(z) in case a perfect channel isavailable.

[0135] The signal R(z) can be derived by observing the reception of asingle data pulse, inserted in T(z). In that case, we do not need tostore the R(z) coefficients, but rather they can be derived at run time.

[0136] Receiver Requirements in the Presence of Carrier Frequency andPhase Errors

[0137] The received signal in the presence of carrier frequency andphase errors is given by: $\begin{matrix}{{S_{d}(z)} = {{S_{c}\left( {z \cdot ^{{- j} \cdot 2 \cdot \pi \cdot \frac{\gamma}{f_{s}}}} \right)} \cdot ^{{- j} \cdot \varphi}}} & (9)\end{matrix}$

[0138] Substituting for the transmitted signal S_(t)(z), and afterfiltering by the receiver filter R(z) we find: $\begin{matrix}{{S_{d}(z)} = {{S_{t}\left( {z^{m} \cdot ^{{- j} \cdot m \cdot 2 \cdot \pi \cdot \frac{\gamma}{f_{s}}}} \right)} \cdot {T\left( {z \cdot ^{{- j} \cdot 2 \cdot \pi \cdot \frac{\gamma}{f_{s}}}} \right)} \cdot {R(z)} \cdot ^{{- j} \cdot \varphi}}} & (10)\end{matrix}$

[0139] ISI-free operation can be obtained if $\begin{matrix}{{{R(z)} = {{T\left( \left( {z \cdot ^{{- j} \cdot 2 \cdot \pi \cdot \frac{\gamma}{f_{s}}}} \right)^{- 1} \right)} \cdot ^{{+ j} \cdot \varphi}}}{{So},}} & (11) \\{{R(z)} = {{- {T\left( {z^{- 1} \cdot ^{j \cdot 2 \cdot \pi \cdot \frac{\gamma}{f_{s}}}} \right)}} \cdot ^{{+ j} \cdot \varphi}}} & (12)\end{matrix}$

[0140] We see that the required matched filter R(z) now is a root-raisedcosine, but whose coefficients are rotated over a constant factor +φ,and a linearly increasing factor −γ.t.

[0141] In contemporary modems, the frequency and phase correction isoften implemented in a separate block, and the special complex filterR(z) is not needed. The receiver presented here however will make use ofthis filter: by observing an incoming data pulse that has a certaincarrier frequency and/or phase error, and taking the complex conjugateand time reverse of the observation, exactly the same formula isobtained.

[0142] Thus, the receiver filter R(z) can be constructed out of theobservation of a single data pulse transmitted through the filter T(z).This will compensate the phase error and the immediate frequency error.A sustained frequency error will however require the use of an adaptivestructure in the modem.

[0143] Receiver Requirements in the Presence of Delay and AttenuationErrors

[0144] The received signal in the presence of delay and attenuationerrors is given by:

S _(d)(z)=K.Z(S _(c)(jω).e ^(−jωn) ₁)).z ^(−n) ₁  (14)

[0145] where K is the attenuation factor of the system, n₁ is theintegral time delay measured in samples of the discrete-time system, andn₂ is the fractional delay. Note that a frequency domain formulation wasused to express the fractional time delay since there is no elegant wayto write this in the Z-domain. We will call the time-shifted waveformS′_(c)(z).

[0146] Substituting for the transmitted signal S_(t)(z), and afterfiltering by the receiver filter R(z) we find:

S _(d)(z)=K.S′ _(t)(z ^(m)).T′(z).R(z).z ^(−n) ₁  (15)

[0147] where T′(z) and S′_(t)(z^(m)) are time-shifted waveforms of thetransmitter.

[0148] The detection problem can be simplified by the following twoobservations:

[0149] We can eliminate the z^(−n) ₁ factor provided we introduce in thereceiver the ability to perform burst start-detection and alignmentitself. The ISI-free operation now is obtained if R(z). T′(z) is anISI-free pulse.

[0150] The gain distortion factor K can be eliminated by including AGC(automatic gain control) circuitry into the receiver. This circuit willautomatically apply a gain correction K¹ onto the received signal.

[0151] Assuming these two circuits are included in the receiver (beforethe filter R(z)), we now find

S _(d)(z)=S′ _(t)(z ^(m)).T(z).R(z)  (16)

[0152] The ISI-free condition now becomes:

R(z)=T′*(z ⁻¹⁾  (17)

[0153] If the transmitter filter T(z) is a root raised cosine filter,then we see that the receiver filter R(z) must be a time-shifted rootraised cosine filter, with a time shift which is opposite to that of thetransmitter shapes. By taking the observation of a transmitted datapulse and time-reversing it, we obtain the required matched filter.

[0154] In contemporary modems, the delay correction is often implementedin a separate block, and the time shifting of filter R(z) is not needed.The receiver of the present preferred embodiment however will make useof this filter.

[0155] Receiver Requirements in the Presence of Group Delay Distortion

[0156] Next, we seek the conditions for ISI-free operation in case of ageneral channel filter C(z).

[0157] In this case, the received signal is:

S _(d)(z)=S _(c)(z)C(z)=S _(t)(z ^(m))T(z)C(z)R(z)  (18)

[0158] and for ISI free operation we need

R(z)=T*(z ⁻¹).C ⁻¹(z)  (19)

[0159] This successful reception requires spectral inversion of thechannel characteristic.

[0160] As before, T*(z⁻¹) is a root-raised-cosine pulse.

[0161] In case of a channel with an all pass characteristic, the inverseof C(z) is derived by making use of the all-pass property of thechannel. Writing down the frequency response of the all-pass channelC(z), we find:

C(jω)=|1|.e ^(j.φ(ω))  (20)

[0162] from which the inverse is found to be:

C ⁻¹(jω)=|1|.e ^(−j.φ(ω)) =C*(jω)  (21)

[0163] We can also substitute jω by e^(jω), which is the definition ofthe Z-transform variable, and verify that C⁻¹(z)=C*(z⁻¹).

[0164] ISI-free operation is obtained by observing an incoming datapulse T(z)C(z), and taking the complex-conjugate and time reverse of theobservation, and forming the required matched filter.

[0165] Noise Susceptibility by Observing a Data Pulse

[0166] In the preceding derivations, it was found that the requiredmatched filter always can be derived through the observation of a singletransmitted data pulse. When the channel is corrupted by noise, theobservation will also be degraded. As a result, the receiver filtercoefficients in R(z) will contain the same noise power level as thenoise power level on the channel.

[0167] The method of pulse observation therefore introduces twice theamount of noise over that of the theoretical method. In terms of modemperformance, this yields a loss of 3 dB, when expressed in power levels.For the upstream CATV channel, which is a shielded environment, thisdegradation is acceptable due to the big advantages of receiversynchronization that are gained out of the method.

First Preferred Embodiment of the Invention

[0168] As a summary, the receiver has the following requirements:

[0169] To construct a matched filter for the distortions, it needs to beable to perform a spectral inversion of the received single data pulse.This data pulse defines essentially the content of the preamble.

[0170] It needs to be able to do signal burst alignment, which involvesdetection of the start (and end) of the burst signal. It needs to detectthe single data pulse necessary to derive T*(z⁻¹)C*(z⁻¹) and also thestart of actual user data.

[0171] It needs to be able to estimate the gain of the signal.

[0172] A receiver constructed with these properties will be able to copewith the typical upstream CATV signal distortions (delay, attenuation,group delay distortion) for the case of quadrature modulation (such asQPSK or QAM16).

[0173] In the following, we will translate these requirements into areceiver architecture.

[0174] The requirements derived above lead to a generic receiverarchitecture and method of operation. In this section, this will bediscussed. We include the definitions of signals to transmit, as well asthe layout of the generic receiver for these type of signals.

[0175] Burst Format

[0176] The burst format of the transmitted signal is presented in FIG.9.

[0177] Two parts are discerned: a preamble (25) and a payload (26). Bothwere defined above. The preamble is identical on I and Q branches.During the payload, the I and Q branches of the modulation can differ.

[0178] The preamble has two parts: a training part (27) and an alignmentpart (28). The training is meant to derive estimates for gain, delay andgroup delay distortion and to derive the required receiver filter R(z).The alignment part is intended to perform burst alignment, and to detectthe transition from preamble to payload. It is to be noted that inspecific embodiments of the invention several training parts may precedethe alignment part.

[0179] During the preamble (25), the maximum power level is transmitted.For any estimation derived from observing the preamble, this ensuresthat noise sensitivity is minimized. In the preferred embodiment, thepreamble can contain 7 symbols of the upper right constellation symbol,followed by 7 symbols of the lower left constellation symbol.

[0180] Next 2 more upper right constellation symbols are sent followedby 1 lower left constellation symbol. This preamble is identical forboth QPSK and QAM-16 type symbol constellations.

[0181] The preamble is also identical for all types of upstream CATVchannels. Once past the preamble, the receiver is synchronized and datadetection on the payload starts.

[0182] The length of the payload can either be an algorithm parameter orvariable, corresponding to the fixed length mode, respectively the autodetect mode.

[0183] In the fixed length mode, a parameter determines the number ofpayload symbols. After the start of a burst is detected by the receiver,the received number of payload symbols is counted. After theparametrised amount of payload symbols is received, the receptionalgorithm terminates the burst reception.

[0184] In the auto detect mode, both the start and the end of the burstare detected by the reception algorithm autonomously. Hence, the payloadmay have a variable length.

[0185] Generic Receiver Architecture

[0186] The burst signal as defined above can be processed by the genericreceiver architecture shown in FIG. 10.

[0187] The leftmost side of the figure shows the burst signal comprisinga training (27) and alignment part (28) (both in the preamble), and apayload part (26). This signal is fed to a DFFFSE (Downsampling FeedForward Fractionally Spaced Equalizer) block (30) which has fractionallyspaced feed forward equalizer (31) and a decision feedback equalizerwhich receives the output from the FSE (31). The training part of theburst signal is used to extract an estimate of the channel impulseresponse. This estimate is inverted in order to obtain the coefficientsof a fractionally spaced feed-forward (FSE)(31) equalizer.

[0188] In order to extract the training part from the burst signal, acourse alignment of the burst signal is completed. This is obtained bydetecting the power level of the signal processed by the equalizer (31).The inversion required for various cases of channel distortion wasdiscussed in the previous section.

[0189] Next, the precise burst alignment is done using the alignmentpart of the burst signal. By processing the alignment signal in thetiming block, the required downsampling phase for the DFFFSE (30) isfound.

[0190] The resulting signal obtained out of the DFFFSE (30) is free oflinear channel distortions, including phase and amplitude distortions.

[0191] It can now be fed into further receiver parts. In the nextsection, a specific embodiment of the receiver (9) will be presented.More specifically, the DFSFFE will be refined to a MFE structure(matched filter equalizer). Also, all structures surrounding the DFSFFEwill be shown, including both pre-processing parts and post-processingparts.

EXAMPLE

[0192] In this section, the detailed operation of the receiver accordingto a first embodiment of the invention is discussed. First the shape ofthe burst signals to transmit is defined. Next, an overall layout of thereceiver will be given. Following this, the operation of each part isdiscussed.

[0193] Receiver Architecture

[0194] The receiver layout is shown in FIG. 11. It comprises 5subsequent processing stages, each of which have a tree letter acronym.The functionality of each block is briefly discussed below. Since theseare digital processors in the present invention, the sample rate at theentry and exit of each block is also indicated. The sample frequency ofthe system is 10 MHz, while the symbol frequency is 2.5 MHz. For QAM16modulation this yields 10 Mbit/s. We will use the symbol f_(p) to denotethe sample frequency. The sample frequency in the figure is indicated asa number of samples per symbol (sps).

[0195] The DOC block performs down-conversion of the received signal atpass-band and returns a baseband signal as a separate I and Q branch.The input sample rate is 4 samples per symbol, and the carrier frequencyis assumed to be located at one fourth of the sample frequency. If theupstream communications carrier is located at a different carrierfrequency then 2.5 MHz (10 MHz/4), then a pre-processing step must beimplemented that will perform frequency translation from the desiredcarrier frequency to the 2.5 MHz carrier. The output of the DOC isdown-sampled such that it returns a signal containing 2 samples persymbol.

[0196] The AGD block performs activity detection and automatic gaincontrol. It monitors the incoming signal power level. When this powerlevel rises above a pre-set threshold, the presence of a burst isassumed and activity is signalled. Also, the power level of the incomingsignal is adjusted to that of a predefined one, which is the automaticgain control operation.

[0197] The MFE block performs data filtering on the incoming burstsignal. It has programmable coefficients to derive the correct datafilter. The coefficients are extracted from the burst preamble. Thisblock also performs burst alignment and down-samples the2-samples-per-symbol signal down to symbol rate. Once the preamble ispassed, the MFE fixes the filter coefficients and performs datafiltering on the payload signal.

[0198] The LMS block is a small adaptive equalizer that compensatesdrift effects occurring during the burst payload. Also, since it hasequalizer properties, it compensates for the remaining mismatch betweenthe matched filter required for successful communication and the actualfilter coefficients residing in the MFE block. This LMS block is anoption that can be avoided by providing the adaptive means in the MFEblock.

[0199] The DMP block translates the quadrature modulated symbolsavailable at the output of LMS back to data symbols. The data symbolsthen can be further processed by channel decoding and source decoding.

[0200] Down-Conversion: DOC

[0201] The DOC block performs down-conversion of the received burstsignal from pass-band to baseband. It is presented in FIG. 12.

[0202] The incoming signal is down-converted to baseband by mixing itwith a local fixed carrier pulsating at one fourth of the samplefrequency. The mixing operation is implemented by multiplication withsine and cosine values of the carrier, which is particularly attractivefor digital implementation at the given carrier frequency: the sine andcosine values are part of the set (−1, 0, 1).

[0203] Next, the signal is passed to a halfband FIR filter in order toremove the residual image at half of the sample frequency thatoriginates from the down-conversion. A halfband FIR filter is a FIRfilter that cuts off all frequency components above one fourth of thesample frequency. It has the property of having half of the coefficientszero. At the end of the halfband filter, the signal is down-sampled by afactor 2. The spectra at different locations of this block are shown inFIG. 13.

[0204] The halfband FIR requirements are derived by considering FIG. 14,which shows the signal spectrum at the input of the halfband filter. Thereceived signal is modulated with a root-raised-cosine pulse shaping.This shaping has one parameter, the roll-off factor r. This factordetermines how far the modulation extends beyond the minimum requiredbandwidth to allow ISI-free operation. This bandwidth is equal to thesymbol frequency.

[0205] Since the symbol frequency is one fourth of the sample frequency,the signal bandwidth thus is fp/4(1+r). The halfband filter has toremove the image band, located the fp/2, that results from thedown-conversion. The dashed line shows the desired response of thefilter. Besides this response requirement, the halfband FIR also needsto have a flat response inside the signal band, in order not tointroduce amplitude distortions. The required suppression d that isoffered by the roll-off factor is a function of the residual ISI levelthat is observed during data detection in the receiver.

[0206] For contemporary modems, the roll-off factor r resides between0.25 and 0.5. Since the receiver is capable of receiving burst signalswith varying roll-off factor, the halfband filter therefore should bedesigned to pass the widest burst signal (the one with the highestroll-off factor).

[0207] Given these observations, the halfband filter is designed using aminimum-squares Parks-McLellan algorithm, with two bands:

[0208] a pass-band region from DC to fp/8*1.5, with response 1 andrelative ripple 1,

[0209] a stop-band region from fp/2-fp/8*1.5 with response 0 andrelative ripple 1.

[0210] A 20-tap filter yields suppression d=−40 dB, which was foundsufficient for QPSK/QAM16 constellations at the projected noise levels.

[0211] Activity Detection and Gain Setting: AGD

[0212] After the DOC block a baseband signal at 2 samples per symbol isavailable. The baseband signal contains an I branch and a Q branch.

[0213] In the AGD block, this signal is observed to detect the start ofan incoming burst signal. Also, the power level is estimated whenactivity is detected. Using this estimation, a gain correction isapplied.

[0214] The operation of the AGD block is shown in FIG. 15.

[0215] The sampled data signal that is shown is the first part of thepreamble out of the burst signal. The AGD continuously monitors theaverage power level by evaluating the amplitude for each I and Q sample,and then averaging this over 4 signal amplitude estimations.

[0216] This averaging reduces the noise sensitivity. As soon as theaverage exceeds a predefined threshold, the presence of a signal burstis assumed, and activity is signalled to the other blocks in the system.

[0217] Also, the activity turn-on event is used to set the gaincorrection. The gain correction circuit is implemented by means of amultiplication with a constant. The constant is dependent on the on thesignal level observed during the preamble, and is obtained by dividing areference value by the observed value. This way, the signal is scaled tothe correct reference.

[0218] The AGD operation is shown graphically in FIG. 16. The upper partshows the signal burst, while the lower part shows the output of theaveraging as a function of time. As the burst starts, the observedsignal level increases. As the average increases above theturn-on-threshold, activity-on is signalled. The gain programmingcircuitry then delay this event by an amount D, before deciding on theactual signal level. As the burst ends, the average drops again, and assoon as it decreases below the turnoff threshold, activity-off issignalled.

[0219] Matched Filter Evaluation: MFE

[0220] The signal that leaves the AGD block has no gain error left.Also, two events are available (activity_on and activity_off), that flagthe start and end of the signal burst.

[0221] The next task is now to perform the data filtering. This is donein the MFE block. The general operation of this block is as follows:during the preamble, the MFE coefficients are determined. Next, burstalignment is performed, and the transition to the payload is detected.During the payload, the MFE coefficients are used to perform datafiltering.

[0222] First, it will be explained how the MFE coefficients aredetermined. As was demonstrated in the receiver requirements derivation,the coefficients are a time-reversed complex-conjugate of a single datapulse.

[0223] Thus, given a sampled impulse response of the channel K(z), theMFE coefficients need to be programmed with K*(z⁻¹). In the presentedburst preamble, there is however no isolated data-pulse. Rather, thechannel impulse response is extracted from the preamble by means of amathematical transformation, as shown in FIG. 17. The impulse responseinformation is extracted from the first down-going transition in thepreamble (indicated in the dashed square in the figure). The step (B) onthe I and Q branch is differentiated with respect to time in order toobtain an impulse response. Since this is a negative impulse occurringon both the I and Q branch, another transformation is needed to return apositive, real unit impulse. To do this, we rotate the observeddifferentiated response over 135 degrees and also scale the signal levelto unity. This simple two-step process thus allows to obtain the channelimpulse response out of the step function encoded in the preamble. Thismethod works for impulse responses that are shorter then the width of alimited window around (B). For longer impulse responses, other events inthe preamble (like (A) or (C)) will show additional influence and causecoefficient programming errors. In turn, these programming errors causesISI. Such a longer impulse response occurs for instance due to thepresence of high amounts of group delay distortion.

[0224] Next, the MFE is discussed. FIG. 18 shows the layout of the MFEblock. It consists of:

[0225] A data delay line. This is needed to perform the data filteringoperation. Given a data filter dimensioning of N coefficients, we needthe data delay line to contain N+2 samples. The two extra samples areneeded to perform successful differentiation. The data delay line hasboth an I and Q branch.

[0226] A differentiator+rotator. This algorithm performs the operationpresented in FIG. 17. Below this operation is expressed mathematically.

[0227] Given the data samples d[N+2], the (discrete) differentiation isevaluated as:

d′[i]=d[i+2]−d[i]  (22)

[0228] As is seen, the differences are taken with a step of two. This isbecause the estimation is done on an over-sampled signal (two samplesper symbol).

[0229] Next, the data sequence is rotated. The rotation over 135 degreesis an easy linear combination. Given a complex number i+jq, the rotatedversion is given by:

i′=−i/sqrt(2)−q/sqrt(2)  (23)

q′=i/sqrt(2)−q/sqrt(2)  (24)

[0230] The normalization factor to obtain a unit impulse is determinedas follows: multiplying the differentiated values with a factor1/6.1/sqrt(2) will reduce the pulse height on Q and I branches to1/sqrt(2). The effective pulse length then is equal to the required 1.

[0231] Bringing the differentiation, rotation and scaling operationtogether the required operation of the differentiator/rotator block isfound:

c _(i) [i]={fraction (1/12)}*(d _(i) [i+2]−d _(i) [i])+{fraction(1/12)}+(d _(q) [i+2]−d _(q) [i])  (25)

c _(q) [i]={fraction (1/12)}*(d _(i) [i+2]−d _(i) [i])+{fraction(1/12)}+(d _(q) [i+2]−d _(q) [i])  (25)

[0232] With respect to digital hardware implementation, we see that onlyadder, subtractor and shift operators are necessary.

[0233] A coefficient store to hold the coefficients c[N] after they areevaluated. The coefficient store is updated a well defined time afterthe activity detection event.

[0234] A convolution block, that performs the convolution between d[ ]and c[ ] during payload reception, thereby performing data filtering.

[0235] A correlator/decimeter, that performs the burst alignment. Duringnormal operation, it will down-\sample the input data stream from twosamples per symbol to 1 sample per symbol. The correlator/decimeter andthe operation of it is discussed next.

[0236] The correlator/decimeter operation is shown in FIG. 19. The inputof the correlator is taken from the data filtering (the convolutionoperation). The input signal will be a filtered burst signal. Since thefirst part of the burst signal is used for coefficient extraction, itwill not appear at the convolution output. Therefore it is indicatedwith dashed lines.

[0237] The correlator/decimeter starts as soon as the data filtercoefficients are found. It will first seek the start of the burst byperforming burst alignment. To do this, it correlates the samples comingout of the convolution operation with a special data pattern, equal tothe last part of the preamble. When the correlation value exceeds apredefined parameter, alignment is assumed, and the downsampling canstart. The correlation maximum always occurs at the same unique samplein the burst (since the correlated signals contains no attenuation orgroup delay distortion effects). Therefore, the correct downsamplingphase that goes from 2 samples per symbols to 1 sample per symbol, isuniquely related to the moment of burst alignment.

[0238] Tracking: LMS

[0239] The next part of the receiver is a small adaptive equalizer.Because it is small, it can fairly fast adapt to changes in the channel.It is intended to remove linear distortions that affect the payload butcannot be corrected by the MFE. These include carrier phase andfrequency errors induced by the transmitter and receiver front-ends, andamplitude distortions induced by the channel.

[0240] At 1 sample per symbol, the LMS equalizer is symbol spaced. Thecoefficient adaptations are steered by means of decision feedback. Sincea receiver decision can always be wrong, the adaptations can potentiallydiverge in case many subsequent decision errors occur. This is a commonproblem in adaptive equalizers; and it is most acute at the moment ofcoefficient initialization.

[0241] In the presented block, the coefficient initialization is howevertrivial: just after MFE programming, the derived matched filterperfectly matches delay errors, and carrier frequency and phase errors.Therefore the required initial LMS coefficients are simply a diracimpulse.

[0242] For the equalizer, 3 or 2 feed-forward taps and 2 or 1 feedbacktaps are chosen, as shown in FIG. 20. It is also possible to useimplementations with another number of feed-forward and feedback taps.With this receiver concept, it is possible to restrict the number oftaps. Note that if the number of taps is increased, the convergence timeof the equalizer will increase too. The power of this invention is thepossibility for fast convergence of the described method, taking intoaccount that the initial situation is almost ideal. Possible candidatesfor number of taps are 2 to 5, or higher.

[0243] The cursor symbol is installed at the tap just before thedecision device. For simplicity, only part of the complete structure isshown. The figure shows all filter taps, the update logic forcoefficient 0 and the accumulation logic for coefficient/tap 3. Thecomplete filter has both of these for each tap and coefficient.

[0244] The equalizing process is programmed by the following parameters:

[0245] QPSK or QAM16 demodulation requires a different decision deviceto be enabled.

[0246] The LMS step size, the factor K in the figure, is made variable.A large step size allows the equalizer to track fast variations, butresults in a greater residual error. A small step minimizes the error,but limits the tracking speed. The variable (programmable) step allowsto set the correct tracking speed for a given environment. A valuesmaller than 0.04 was found appropriate for typical upstream CATVchannels. This low value can be chosen because at the start of thepayload the MFE matches perfectly all channel imperfections.

[0247] Finally, the start of the payload must be signalled in order toproperly initialize the coefficients. The start-of-payload event isgenerated out of the MFE block as a result of burst alignment.

[0248] Symbol-to-Data Translation: DMP

[0249] After data filtering (MFE) and equalization (LMS), the outputsymbols should ideally fall exactly on top of the original transmittedsymbols. In reality, this is never the case. Due to noise effects, whichcannot be compensated, and residual ISI effects, the LMS output symbolscan differ slightly from the original transmitted symbols.

[0250] To retrieve the original symbols, and translate these to the userdata inserted at the transmitter side, the DMP block is used. This blockobserves an LMS output constellation, hard-limits it, and translates thesymbol back to user data values.

[0251] This process is shown in FIG. 21 for the case of a QAM-16constellation. Each constellation point is located at the center of abin. The edges of these bins are the decision levels. An LMS outputsymbol (indicated with an X-cross in the Figure) is compared to thesedecision levels to select a bin in which the actual symbol shouldreside. Once this symbol is decided, it is translated back to a userdata value.

[0252] Because of the hard-limiting, the receiver can make a wrongdecision. Once the LMS output symbol moves across a decision boundary,the DMP block will take another decision symbol. Thus, in case the LMSoutput symbols strongly deviate from the expected positions, it islikely that the wrong symbols will be decided. In that case, atransmission error occurs.

Second Preferred Embodiment of the Invention

[0253] The receiver architecture is now mapped to a preferredimplementation. A burst-mode receiver is conceived such that thehardware initialization time between the reception of two bursts is assmall as possible. This will allow a minimal burst spacing.

[0254] The algorithms presented up to now are of the discrete-time type,for which digital hardware is an ideal container. Only the initialtuning function, that translates the modulation carrier from thetransmission frequency to symbol frequency, needs processing by analogparts.

[0255] The receiver system, shown in FIG. 22, consists of:

[0256] An analog frequency conversion unit

[0257] An analog-to-digital converter

[0258] A digital burst demodulator

[0259] Since the symbol timing correction is performed inside of thedigital burst demodulator, the analog-to-digital converter can beclocked at a fixed frequency.

[0260] The following description focuses on the implementation of thedigital demodulation algorithms. In particular, the mechanism thatenables minimal interburst hardware initialization time will bediscussed.

[0261] The receiver algorithms discussed before are synchronized by thepresence of data. Coefficient programming of coefficients in the MFE,for example, is triggered by data detection in the AGD block. Next, oncethe MFE produces valid output data, it triggers the LMS to start dataequalization. It is easily seen that, in this chain of algorithms,synchronization is done between neighboring algorithms. The passing ofdata itself is the synchronization act.

[0262] Each algorithm is executed by a dedicated digital hardwaremachine, comprising a local controller and a data path. The data pathexecutes the data processing operations inside the algorithm, while thelocal controller performs operation sequencing, and algorithmsynchronization.

[0263] In the case of the second preferred embodiment, the data-drivenarchitecture is particularly useful because of the absence of feedbackloops over the chain of algorithms. In that case, all data processingparallelism available through the use of multiple local controllers isobtained. Also, the correct system operation is guaranteed from thecorrect operation of the individual blocks.

[0264] In contrast, when algorithm feedback loops are present, dataprocessing parallelism can be restricted by critical path requirementsof the feedback loops. As an example of this, it can be verified that inthe following pseudocode snippet, process 1 and process 2 cannot everexecute in parallel (since process 2 need value v1, which is handed overdirectly from process 1).

[0265] input1=previous_output2;

[0266] process 1 (input1,v1)

[0267] accept input1;

[0268] do processing;

[0269] send v1;

[0270] process 2 (v1, output2)

[0271] accept v1;

[0272] do processing;

[0273] send output2;

[0274] previous_output2=output2;

[0275] A second drawback of feedback loops in data driven architecturesis that the correct system operation is not guaranteed from the correctoperation of individual components alone.

[0276] We conclude the preferred embodiment of the loosely coupledreceiver algorithm in the invention also are best implemented in adata-driven feed-forward architecture.

[0277]FIG. 23 shows the internals of the digital burst demodulator. Thealgorithmic chain that was presented above is recognized. Apart from thefirst block (DOC), in the preferred embodiment, all blocks consist ofboth a finite state machine (FSM) and a datapath (DP). Such blocks havemore than one mode of operation, and behave differently when a burst isbeing received than when not. Datapaths are constructed from bitparalleldigital hardware such as registers, adders, multipliers and shifters.Finite state machines are constructed from digital hardware includingrandom logic and registers.

[0278] The chain of blocks includes:

[0279] A down-conversion block DOC, for conversion from pass-band tobaseband.

[0280] The AGD block, for activity detection and gain control.

[0281] The MFE block, for matched filtering, timing correction and delaydistortion correction.

[0282] The LMS block, for adaptive equalization of the channel.

[0283] The DMP block, for symbol detection.

[0284] The wide arrows between the blocks show the flow of data. Thedata, consisting of samples of the burst signal, is passed from datapathto datapath.

[0285] The thin arrows show the flow of control. Arrows directed to anFSM block indicate an event being signalled to the control programrunning in the FSM. Arrows directed from an FSM indicate a particularmode of operation being selected. When such a mode of operation ispassed to a datapath block, it is interpreted as an instruction by thisdatapath. When it is passed to another FSM, it signals a particularevent to this other FSM.

[0286] It is seen that the flow of control is free of feedback loops.More specifically, the mode of operation is controlled from left toright. The signals c1 through c7, and a1 through a3 have the followingfunction:

[0287] c1 signals that the turn-on-threshold of the AGC is exceeded.This is interpreted as the start-of-burst condition by the AGD FSM. Itis obtained by observing the power of the incoming burst signal, as wasdiscussed before.

[0288] The AGD/FSM observes c1 and transmits c2 the MFE/FSM. c2 is adelayed version of c1. It is transmitted when the training sequence hasrippled into the MFE/DP block.

[0289] The instruction on a1, sent to the MFE/DP block, selects when theMFE coefficients are to be programmed. The MFE/DP also detects the burstalignment pattern at the end of the preamble. In that case, an event onc5 is passed to the MFE/FSM. Based on this event, the MFE/FSM can thenselect the correct downsampling phase for the MFE/DP.

[0290] The event on c3, passed from MFE/FSM to LMS/FSM, indicates whenthe MFE coefficients have been obtained, burst alignment has been done,and a valid signal is passed to LMS/DP. This event is translated into aninstruction on a2, that will set the coefficients equalizer in LMS/DP totheir initial state.

[0291] When the LMS/DP block produces the first sample of valid data,the LMS/FSM signals this to DMP/FSM through c4. DMP/FSM starts thedetection of data symbols on DMP/DP through a3. The symbols are alsocounted, and when the preselected burst count is reached, the start ofdata symbol detection signal a3 is released and the end-of-frame isindicated through c6 and subsequently through c7.

[0292] In case of the auto detect mode, the symbols are not counted inorder to terminate the burst reception, but rather an end-of-burstcondition is evaluated by observing the power of the incoming burstsignal. The condition evaluates to true when the turnoff-threshold isexceeded.

[0293] Because the flow of control is essentially feed-forward, thealgorithm becomes independent of the latency of individual blocks. Itis, for example, possible that the AGD and MFE are decoding one burstwhen the LMS and DMP are still processing the previous one. Theinterburst spacing is therefore only needed to allow the AGD to detect aburst gap.

[0294] Performance of the Receiver According to the Preferred Embodimentof the Invention

[0295] In this section, the performance of the receiver is examined. Thefollowing transmission distortions are considered:

[0296] Noise with a gaussian distribution is added to the burst signalbefore reception starts. The noise power level is varied with relationto the burst signal power level to see how sensitive the receiver iswith relation to noise influences.

[0297] Group delay distortion is introduced on the burst signal beforereception.

[0298] Symbol timing drift and carrier phase/frequency drift isintroduced during the payload to test the adaptiveness of the receiver.

[0299] The power level of the burst signal is varied to see theoperation of the AGD.

[0300] Amplitude distortions are introduced on the burst signal beforereception.

[0301] In order to judge the performance in an objective way, a qualitymeasure of the reception has to be used.

[0302] Two quality measures are referenced here:

[0303] The number of symbol errors occurring in the transmission of aburst. This translates to a symbol error rate, which is the ratio of thenumber of symbol errors to the total number of transmitted symbols.

[0304] The Errored Vectored Measure (EVM) constellation quality measure.The EVM constellation quality measure is defined as follows:$\begin{matrix}{{EVM} = \sqrt{\frac{1}{N}{\sum\limits_{N}^{\quad}\left( {\left( {I - I_{r}} \right)^{2} + \left( {Q - Q_{r}} \right)^{2}} \right)}}} & (27)\end{matrix}$

[0305] where N equals the number of symbols that are included in thequality estimate, I and Q the symbol value of the received symbol, I_(r)and Q_(r) the symbol decided by the DMP block. Since this formula stillis dependent on the absolute size of the constellation, a normalizationconvention is used: the outer constellation symbols are assumed to belocated at 1. The EVM quality measure is dimensionless and expressed asa percentile value. For judging the reception of a burst signal, we usethe EVM performance figure as obtained from the symbols contained in thesignal burst (Thus N is equal to the number of symbols in the payload).

[0306] The EVM figure has no direct relation to symbol errors. This isbecause the EVM figure makes abstraction of the distribution of thereceived symbols around a desired constellation point.

[0307] If we make an assumption about this distribution, we can howeverobtain an approximate relation. Table 2 shows this relation for the caseof a gaussian distribution (which occurs when only gaussian noise ispresent in the channel). TABLE 2 Relation Symbol Error rate and EVM fora gaussian distribution EVM (%) SER 7 3.10⁻⁶ 8 5.10⁻⁵ 9 3.10⁻⁴

[0308] The target symbol error rate for the receiver is a 5.10⁻⁵ symbolerror rate under realistic channel conditions. In that case, theachieved bit error rate including channel coding obtained by thereceiver is comparable to many of the contemporary high-speed cell-basedcommunication systems (bit error rate=10⁻¹⁰).

[0309] Noise Performance

[0310] The receiver was tested in a Monte Carlo set-up, where very longbursts were transmitted until 10 symbol errors were observed. Accordingto statistics theory, an observation of 10 events (symbol errors)guarantees a certainty of 90 percent that the actual value is less then50 percent wrong. This is generally considered as a good estimate.

[0311]FIG. 24 shows the receiver performance at different noise levelsin the channel. The continuous lines are the ones obtained bytheoretical considerations of achievable receiver performance. To verifythese, a model of this perfect receiver was constructed, and a MonteCarlo simulation was set-up. The circles in the Figure show simulatedworking points.

[0312] Next, the proposed receiver for burst-mode-transmission wassimulated. The results for QPSK and QAM16 modulated bursts are indicatedwith diamond and cross shapes respectively.

[0313] Group Delay Distortion Performance

[0314] By simulating different amounts of group delay distortion on areceived burst signal, it is estimated how quickly the EVM performanceof the receiver degrades (due to residual ISI effects). The figures intable 3 show the obtained performance for a QAM16 with a symbol periodof 400 ns (symbol rate 2.5 Msym/s), and 10 filter taps (10 I and 10 Qcoefficients) in the MFE block. To demonstrate the power of thisequalization concept, a similar simulation was set-up on a receiver thatuses no adaptive, but fixed data filter coefficients. This is shown inthe third column of the table. This clearly demonstrates the advantagesof using the MFE. TABLE 3 Achieved EVM for various group delaydistortions Group Delay Distortion (ns) EVM EVM wo MFE 0 1.4 1.4 65 2.03.3 325 3.0 16.0 650 5.5 more than 20

[0315] Symbol Clock Drift Performance

[0316] The clock of the digital receiver is chosen to be a multiple ofthe transmitted symbol \rate. Due to component tolerances of this clock,the actual symbol clock can slightly vary. This is usually expressed asa ppm value, which indicates how big the misalignment is afterobservation of one million symbols.

[0317] Since the MFE block coefficients are fixed after reception of thepreamble, the required matched filter for reception therefore graduallydiverges from the matched filter obtained during the preamble.

[0318] To test the sensitivity of the receiver to this, the followingtest was made. A payload length of 424 data bits and 48 channel codingbits was assumed. This message corresponds to 118 QAM symbols (4bits/symbol) or 236 QPSK symbols (2 bits/symbol).

[0319] A clock deviation of 80 ppm was assumed. During the reception ofone burst, the divergence of the symbol clock therefore diverges lessthen 80.10⁻⁶*118=0.00944 symbols for the QAM16 case, and less then80.10⁻⁶* 236=0.0188 symbols for the QPSK case.

[0320] In both case, the deviation is considered to be negligible (EVMdeviations in the order of one tenth %). We therefore conclude that thedrift on the symbol clock in a burst-mode transmission system has noobservable effect on performance.

[0321] Carrier Phase Drift Performance

[0322] Besides drift on the symbol clock, there is also drift on thecarrier frequency and phase because of manufacturing imperfections. Alsothis figure is expressed as a ppm value, and is interpreted as thedeviation of the carrier frequency with respect to its nominal value.The carrier frequency at the receiver entrance is equal to the symbolrate. Therefore one revolution of the carrier corresponds to one symbolperiod. Similarly, a drift of e.g. 10 ppm means that the carrier phasedeviates 10 revolutions in one million. A deviation of one revolutioncorresponds to the QAM16 or QPSK constellation rotating 360 degrees.

[0323] The sensitivity on carrier phase drift is higher than the drifton symbol timing, because the ppm tolerance specification applies to thecarrier frequency. The carrier phase drift is the integrated effect ofthis tolerance.

[0324] To counter effects of Carrier Phase Drift, the receiver uses theLMS block. In table 4, the receiver performance is tabulated for varioustolerances on the carrier frequency clock. The first column shows theperformance with an active LMS, the second one the performance when thecoefficient adaptations in the LMS block are disabled. The figuresjustify the need for this block in the receiver architecture. TABLE 4Achieved EVM for various carrier phase drift Carrier Frequency Tolerance(ppm) EVM EVM wo LMS 10 2.0 3.0 30 3.8 7.9 50 6.5 13.5

[0325] Gain Control Performance

[0326] Each burst that arrives at the receiver has travelled through adifferent signal path, and hence has also a different gain. The AGDblock corrects gain errors on the signal. This block also performsactivity detection, and this is done by fixed threshold detection.

[0327] A potential problem is that the activity detect point (relativeto the position of the burst) becomes dependent of the gain. This isexplained in FIG. 25, which shows the nominal case.

[0328] The start of a burst signal is shown as seen at the output of theDOC block (two samples per symbol).

[0329] Around sample 38, the signal power on the I and Q branch rises,and crosses the on-threshold value. An activity detect is signalled,which will program the MFE coefficients. The position of the MFEcoefficients in the burst is therefore related to the moment at whichactivity is detected. In case of a gain error, this moment varies (sincethe threshold level is fixed).

[0330] The effect of varying burst gain and holding the threshold-onlevel fixed is shown in FIG. 26. 10 taps are assumed in the MFEstructure. The ideal case occurs at relative (burst) signal amplitude 1.At that moment, the matched filter peak (the center of the matchedfilter impulse response) coincides with the central MFE taps, 4 and 5.When the signal grows or weakens, activity detection is done earlier orlater respectively, with relation to the burst edge. Therefore, thematched filter peak starts also to shift back responsibility forward ofthe MFE matched filter.

[0331] The shifting of the matched waveform in the MFE block is adiscrete process since the MFE contains a limited amount of taps. Thecentral peak shifts one sample at a time, which causes the matchedfilter waveform in the MFE to be aligned alternately on half and fullsymbol boundaries. The half symbol boundary case (for example, ⅚ and ¾)is clearly inferior to full symbol boundary alignments.

[0332] The Figure indicates that there is approximately 10 dB of dynamicrange before the matched filter waveform starts to shift in the MFEtaps. If we consider EVM=3.5% to be an upper limit, we find well over 30dB of dynamic range.

[0333] Amplitude Distortion Performance

[0334] As a last performance measure, the effect of amplitudedistortions on the receiver is investigated. Although the CATV channelis designed to have a flat response, small amplitude variations willalways occur.

[0335] These are mostly caused by impedance mismatches in the network,which result in signal reflections. In turn, these result in amplitudevariations of the channel response, commonly called amplitude ‘ripple’.The results of a trial experiment are shown in FIG. 27. A model of thedistortion is shown in the upper part of the figure. Three differentchannel profiles are shown. Each of them has a certain amount of rippleon them. In this example, the profile has 10 extreme (minima or maxima).

[0336] Below, the modem performance is shown for an amplitude profilewith 5 extreme, and one with 20 extreme. In both cases the LMS block wasenabled and disabled, in order to see the performance difference as afunction of amplitude ripples.

[0337] The architecture with LMS is suited to counter ripple amplitudedistortion, though the compensation power is limited. However, thisproperty is limiting for all modems working by equalizer techniques, aswill be shown next.

[0338] Consider the ripple model of FIG. 28. The upper part shows thetime domain response, while the lower part shows the frequency domainresponse.

[0339] A simple rule of thumb can now be derived by considering therelation of the frequency domain peak-to-peak ripple distance and thetime domain echo separation. A 1 MHz peak-to-peak ripple distance thuscorresponds to an echo 1 microsecond after/before the main time domainshape.

[0340] Using this rule of thumb, we can now count the number ofamplitude ripples in the modulated band to arrive at an estimate of therequired number of taps for a symbol spaced equalizer (such as found inthe LMS block). 20 ripples inside of the signal modulation bandwidthcorrespond to an echo that extends over 20 symbol period. Thus, at least20 taps are needed in the equalizer to remove the ISI induced by thisecho.

[0341] The following is concluded: The more amplitude ripples in themodulation bandwidth, the more taps required for the equalizer thatprovides compensation. On the other hand, a longer equalizer adapts moreslowly, and thus makes tracking more difficult. For this reason theequalizer in the proposed architecture is deliberately chosen short.

What is claimed is:
 1. A telecommunication system with an upstreamcommunication system for communication from a user to a head-end over achannel, said upstream communication system having a head-end receivercomprising a detect unit configured in a feed-forward data-drivenarchitecture.
 2. The telecommunication system as recited in claim 1,wherein said detect unit is adapted to receive a burst-mode signal, saidburst-mode signal having a preamble with at least one training portionand at least one timing alignment portion.
 3. The telecommunicationsystem as recited in claim 2, wherein said at least one training portionis positioned at the beginning of a communication burst and is followedby said at least one timing alignment portion.
 4. The system as recitedin claim 3, wherein said burst-mode signal further comprises a usermessage.
 5. The system as recited in claim 3, wherein said user messagecomprises error correcting codes.
 6. The communications system asrecited in claim 4, said upstream communication system having atransmitter for said user message, said user message being a firstsequence of data, said transmitter comprising a preamble insertion unitconfigured to insert a predetermined sequence of data to said firstsequence of data, said predetermined sequence having said trainingportion at the beginning of the predetermined sequence followed by saidtiming alignment portion.
 7. The communication system as recited inclaim 6, wherein said preamble insertion unit is configured to prependsaid predetermined sequence of data to said first sequence of data. 8.The system as recited in claim 6, wherein the training portion comprisesa step signal.
 9. The system as recited in claim 8, wherein the lengthof said user message is an algorithm parameter.
 10. The system asrecited in claim 8, wherein the length of said user message is variable.